Differential amplifying circuit

ABSTRACT

Disclosed is a differential amplifying circuit including an amplifying circuit, wherein 1) a drain of a sixth transistor is connected to a drain of an eighth transistor, and a drain of a tenth transistor is connected to a drain of a fourth transistor, and 2) a ratio between a total of gate widths of the fourth (or eighth) and tenth (or sixth) transistors (converted per unit gate length, and gate widths that follow are the same) and a gate width of a fifth (or ninth) transistor is nearly proportional to a current ratio between a first (or third) and second (or fourth) current source circuits, the gate width of the fourth (or eighth) transistor being equal to or more than that of the tenth (or sixth) transistor.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is based upon and claims the benefit of priority fromand is a continuation of application Ser. No. 11/618,071 filed on Dec.29, 2006, which is based upon and claims the benefit of priority fromthe prior Japanese Patent Application No. 2006-66639, filed on Mar. 10,2006; the entire contents of both of which are incorporated herein byreference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a differential amplifying circuitsuitable for an integrated circuit.

2. Description of the Related Art

For example, to realize a high-precision pipelined A/D converter, adifferential amplifying circuit with a high differential DC gain(hereinafter expressed as “DC gain”) is needed. One of configurations toobtain a high DC gain in the differential amplifying circuit is aconfiguration including a gain boost amplifying circuit. The DC gain ofthe differential amplifying circuit can be increased by a DC gain of theadded gain boost amplifying circuit. As an example of an A/D converterusing the differential amplifying circuit including the gain boostamplifying circuit, there is one disclosed in the following related art1.

A gain boost amplifying circuit disclosed in this related art has aconfiguration in which a transistor is added to adjust the output commonmode voltage to a vicinity of a median value between reference electricpotentials vdd and vss (hereinafter expressed as “reference potentialsvdd and vss”), and power consumption increases by an amountcorresponding to a current flowing through this transistor.

[Related Art 1] Yun Chiu et al., “A 14-b 12-MS/s CMOS pipeline ADC withover 100-dB SFDR”, IEEE Journal of Solid-State Circuit, United States,IEEE, December 2004, Vol. 39, No. 12, pp. 2139-2151

BRIEF SUMMARY OF THE INVENTION

A differential amplifying circuit according to one aspect of the presentinvention includes: an input stage including a pair of differentialinput terminals and a pair of differential output nodes outputtingdifferential currents according to differential voltages inputted to thepair of differential input terminals; a first intermediate stageincluding a first transistor and a first amplifying circuit, the firsttransistor having a source to which one of the pair of differentialoutput nodes and an input side of the first amplifying circuit areconnected, a gate to which an output side of the first amplifyingcircuit is connected, and a drain being a negative-side current outputnode; a second intermediate stage including a second transistor and asecond amplifying circuit, the second transistor having a source towhich another of the pair of differential output nodes and an input sideof the second amplifying circuit are connected, a gate to which anoutput side of the second amplifying circuit is connected, and a drainbeing a positive-side current output node; and an output stage using thenegative-side current output node and the positive-side current outputnode as a pair of differential input nodes and including a pair ofdifferential output terminals outputting differential voltages accordingto differential currents inputted to the pair of differential inputnodes, wherein the first amplifying circuit includes: a first and secondcurrent source circuits whose one ends are connected to a firstreference potential; a third transistor having a source to which one ofthe differential output nodes in the input stage is connected and a gateto which a bias voltage is applied, and across which a bias currentflows caused by the first current source circuit; a fourth transistorhaving a source connected to a second reference potential, a drain towhich a current is inputted from the third transistor, and a gateconnected to a drain of the third transistor; a fifth transistor havinga gate and a source connected in common with those of the fourthtransistor respectively and a drain to which a current from the secondcurrent source circuit is inputted; and a sixth transistor having a gateand a source connected in common with those of the fourth transistorrespectively, the source of the third transistor being the input of thefirst amplifying circuit, and the output of the first amplifying circuitbeing on a drain side of the fifth transistor, the second amplifyingcircuit includes: a third and fourth current source circuits whose oneends are connected to the first reference potential; a seventhtransistor having a source to which another of the differential outputnodes in the input stage is connected and a gate to which a bias voltageis applied, and across which a bias current flows caused by the thirdcurrent source circuit; an eighth transistor having a source connectedto the second reference potential, a drain to which a current isinputted from the seventh transistor, and a gate connected to a drain ofthe seventh transistor; a ninth transistor having a gate and a sourceconnected in common with those of the eighth transistor respectively anda drain to which a current from the fourth current source circuit isinputted; and a tenth transistor having a gate and a source connected incommon with those of the eighth transistor respectively, the source ofthe seventh transistor being the input of the second amplifying circuit,and the output of the second amplifying circuit being on a drain side ofthe ninth transistor, a drain of the sixth transistor is connected tothe drain of the eighth transistor, and a drain of the tenth transistoris connected to the drain of the fourth transistor, a ratio between atotal of gate widths converted per unit gate length of the fourth andthe tenth transistor and a gate width converted per unit gate length ofthe fifth transistor is nearly proportional to a current ratio betweenthe first current source circuit and the second current source circuit,the gate width converted per unit gate length of the fourth transistorbeing equal to or more than the gate width converted per unit gatelength of the tenth transistor, and a ratio between a total of gatewidths converted per unit gate length of the eighth and the sixthtransistor and a gate width converted per unit gate length of the ninthtransistor is nearly proportional to a current ratio between the thirdcurrent source circuit and the fourth current source circuit, the gatewidth converted per unit gate length of the eighth transistor beingequal to or more than the gate width converted per unit gate length ofthe sixth transistor.

In this differential amplifying circuit, twists are added to the firstand second amplifying circuits included therein. Namely, 1) the drain ofthe sixth transistor is connected to the drain of the eighth transistor,and the drain of the tenth transistor is connected to the drain of thefourth transistor. 2) The ratio between the total of gate widthsconverted per unit gate length of the fourth and the tenth transistorand a gate width converted per unit gate length of the fifth transistoris nearly proportional to the current ratio between the first currentsource circuit and the second current source circuit, the gate widthconverted per unit gate length of the fourth transistor being equal toor more than the gate width converted per unit gate length of the tenthtransistor. 3) The ratio between the total of gate widths converted perunit gate length of the eighth and the sixth transistor and a gate widthconverted per unit gate length of the ninth transistor is nearlyproportional to the current ratio between the third current sourcecircuit and the fourth current source circuit, the gate width convertedper unit gate length of the eighth transistor being equal to or morethan the gate width converted per unit gate length of the sixthtransistor.

Consequently, the parallel output resistance of the fourth and tenthtransistors and the parallel output resistance of the eighth and sixthtransistors rise, which can increase the gains of the first and secondamplifying circuits. Accordingly, as the differential amplifyingcircuit, its DC gain can be increased by an increase in the DC gains ofthe first and second amplifying circuits. Incidentally, a transistor toadjust the output common mode voltage to a vicinity of a median valuebetween reference potentials vdd and vss is unnecessary, which leads toa reduction in power consumption.

Further, a differential amplifying circuit according to another aspectof the present invention includes: an input stage including a pair ofdifferential input terminals and a pair of differential output nodesoutputting differential currents according to differential voltagesinputted to the pair of differential input terminals; a firstintermediate stage including a first transistor and a first amplifyingcircuit, the first transistor having a source to which one of the pairof differential output nodes and an input side of the first amplifyingcircuit are connected, a gate to which an output side of the firstamplifying circuit is connected, and a drain being a negative-sidecurrent output node; a second intermediate stage including a secondtransistor and a second amplifying circuit, the second transistor havinga source to which another of the pair of differential output nodes andan input side of the second amplifying circuit are connected, a gate towhich an output side of the second amplifying circuit is connected, anda drain being a positive-side current output node; and an output stageusing the negative-side current output node and the positive-sidecurrent output node as a pair of differential input nodes and includinga pair of differential output terminals outputting differential voltagesaccording to differential currents inputted to the pair of differentialinput nodes, wherein the first amplifying circuit includes: a first andsecond current source circuits whose one ends are connected to a firstreference potential; a third transistor having a source to which one ofthe differential output nodes in the input stage is connected and a gateto which a bias voltage is applied, and across which a bias currentflows caused by the first current source circuit; a fourth transistorhaving a source connected to a second reference potential, a drain towhich a current is inputted from the third transistor, and a gateconnected to a drain of the third transistor; a fifth transistor havinga gate and a source connected in common with those of the fourthtransistor respectively and a drain to which a current from the secondcurrent source circuit is inputted; an eleventh transistor having asource connected to the drain of the fourth transistor and a gate towhich a bias voltage is applied, and across which a bias current flowscaused by the first current source circuit; a twelfth transistor havinga source connected to the drain of the fifth transistor, and acrosswhich a bias current flows caused by the second current source circuit;and a first sub-amplifying circuit performing amplification with thesource of the eleventh transistor and the source of the twelfthtransistor as bipolar inputs and supplying an output thereof to a gateof the twelfth transistor, the second amplifying circuit includes: athird and fourth current source circuits whose one ends are connected tothe first reference potential; a seventh transistor having a source towhich another of the differential output nodes is connected and a gateto which a bias voltage is applied, and across which a bias currentflows caused by the third current source circuit; an eighth transistorhaving a source connected to the second reference potential, a drain towhich a current is inputted from the seventh transistor, and a gateconnected to a drain of the seventh transistor; a ninth transistorhaving a gate and a source connected in common with those of the eighthtransistor respectively and a drain to which a current from the fourthcurrent source circuit is inputted; a thirteenth transistor having asource connected to the drain of the eighth transistor and a gate towhich a bias voltage is applied, and across which a bias current flowscaused by the third current source; a fourteenth transistor having asource connected to the drain of the ninth transistor, and across whicha bias current flows caused by the third current source; and a secondsub-amplifying circuit performing amplification using the source of thethirteenth transistor and the source of the fourteenth transistor asbipolar inputs and supplying an output thereof to a gate of thefourteenth transistor, the first sub-amplifying circuit of the firstamplifying circuit includes: a fifteenth transistor having a source usedas one of the bipolar inputs and a gate to which a bias voltage isapplied; a sixteenth transistor having a source used as another of thebipolar inputs and a gate to which a bias voltage is applied; aseventeenth transistor having a source connected to the second referencepotential and a gate connected to a drain of the sixteenth transistorand outputting a drain current to the sixteenth transistor, and aneighteenth transistor having a source and a gate connected in commonwith those of the seventeenth transistor respectively and outputting adrain current to the fifteenth transistor, and the second sub-amplifyingcircuit of the second amplifying circuit includes: a nineteenthtransistor having a source used as one of the bipolar inputs and a gateto which a bias voltage is applied; a twentieth transistor having asource used as another of the bipolar inputs and a gate to which a biasvoltage is applied; a twenty-first transistor having a source connectedto the second reference potential and a gate connected to a drain of thetwentieth transistor and outputting a drain current to the twentiethtransistor, and a twenty-second transistor having a source and a gateconnected in common with those of the twenty-first transistorrespectively and outputting a drain current to the nineteenthtransistor.

In this differential amplifying circuit, another twist is added to thefirst and second amplifying circuits included therein. Namely, the firstamplifying circuit includes the first sub-amplifying circuit performingamplification with the source of the eleventh transistor and the sourceof the twelfth transistor as bipolar inputs and supplying an outputthereof to the gate of the twelfth transistor, and the second amplifyingcircuit includes the second sub-amplifying circuit performingamplification using the source of the thirteenth transistor and thesource of the fourteenth transistor as bipolar inputs and supplying anoutput thereof to the gate of the fourteenth transistor.

Here, the first sub-amplifying circuit includes: the fifteenthtransistor having the source used as one of the bipolar inputs and thegate to which the bias voltage is applied; the sixteenth transistorhaving the source used as the other of the bipolar inputs and the gateto which the bias voltage is applied; the seventeenth transistor havingthe source connected to the second reference potential and the gateconnected to the drain of the sixteenth transistor and outputting thedrain current to the sixteenth transistor, and the eighteenth transistorhaving the source and the gate connected in common with those of theseventeenth transistor respectively and outputting the drain current tothe fifteenth transistor. Further, the second sub-amplifying circuitalso has the same configuration.

Consequently, the gains of the first and second amplifying circuits canbe increased. Accordingly, as the differential amplifying circuit, itsDC gain can be increased by an increase in the DC gains of the first andsecond amplifying circuits. Incidentally, a transistor to adjust theoutput common mode voltage to a vicinity of a median value betweenreference potentials vdd and vss is unnecessary, which leads to areduction in power consumption.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS

FIG. 1 is a circuit block diagram showing the schematic configuration ofa differential amplifying circuit according to one embodiment.

FIG. 2 is a circuit diagram showing one example of gain boost amplifyingcircuits G1 and G2 shown in FIG. 1.

FIG. 3 is a circuit diagram showing a general cascode circuit.

FIG. 4 is a circuit diagram showing another example of the gain boostamplifying circuits G1 and G2 shown in FIG. 1.

FIG. 5 is a circuit diagram showing a general active cascode circuit.

FIG. 6 is a circuit diagram showing still another example of the gainboost amplifying circuits G1 and G2 shown in FIG. 1.

FIG. 7 is a circuit diagram showing yet another example of the gainboost amplifying circuits G1 and G2 shown in FIG. 1.

FIG. 8 is a circuit diagram showing one example of sub-gain boostamplifying circuits GS1 and GS2 shown in FIG. 7.

FIG. 9 is a circuit diagram showing yet another example of the gainboost amplifying circuits G1 and G2 shown in FIG. 1.

FIG. 10 is a circuit diagram showing one example of sub-gain boostamplifying circuits GS3 and GS4 shown in FIG. 9.

FIG. 11 is a circuit diagram showing yet another example of the gainboost amplifying circuits G1 and G2 shown in FIG. 1.

FIG. 12 is a circuit diagram showing one example of sub-gain boostamplifying circuits GS1 and GS2 shown in FIG. 11.

FIG. 13 is a circuit diagram showing one example of sub-gain boostamplifying circuits GS3 and GS4 shown in FIG. 11.

FIG. 14 is a circuit diagram showing a configuration of a differentialamplifying circuit as a comparative reference example.

FIG. 15 is a circuit diagram showing one example of gain boostamplifying circuits GJ1 and GJ2 shown in FIG. 14.

FIG. 16 is a block diagram showing a configuration of a pipelined A/Dconverter to which the differential amplifying circuit according to theembodiments can be applied.

FIG. 17 is a circuit diagram showing another example of sub-gain boostamplifying circuits GS1 and GS2 shown in FIG. 7.

FIG. 18 is a circuit diagram showing another example of sub-gain boostamplifying circuits GS3 and GS4 shown in FIG. 9.

DETAILED DESCRIPTION OF THE INVENTION Description of Embodiments

Embodiments of the present invention will be described with reference tothe drawings, but the drawings are presented only for illustrativepurpose and do not limit the invention in any way.

As a form in one aspect, it can be configured that the first amplifyingcircuit further includes: an eleventh transistor having a sourceconnected to the drain of the fourth transistor and a gate to which abias voltage is applied, and across which a bias current flows caused bythe first current source; and a twelfth transistor having a sourceconnected to the drain of the fifth transistor and a gate to which abias voltage is applied, and across which a bias current flows caused bythe second current source, and the output of the first amplifyingcircuit being at a drain of the twelfth transistor, and the secondamplifying circuit further includes: a thirteenth transistor having asource connected to the drain of the eighth transistor and a gate towhich a bias voltage is applied, and across which a bias current flowscaused by the third current source; and a fourteenth transistor having asource connected to the drain of the ninth transistor and a gate towhich a bias voltage is applied, and across which a bias current flowscaused by the fourth current source, and the output of the secondamplifying circuit being at a drain of the fourteenth transistor.

This configuration can increase the gain of the first or the secondamplifying circuit by the product of the transconductances of theeleventh and twelfth transistors and the output resistances thereof andby the product of the transconductances of the thirteenth and fourteenthtransistors and the output resistances thereof. Consequently, as thedifferential amplifying circuit, its gain is further increased.

Further, as a form, it can also be configured that the first amplifyingcircuit further includes a twelfth transistor having a source connectedto the drain of the fifth transistor and a gate to which a bias voltageis applied, and across which a bias current flows caused by the secondcurrent source, and the output of the first amplifying circuit being ata drain of the twelfth transistor, and the second amplifying circuitfurther includes a fourteenth transistor having a source connected tothe drain of the ninth transistor and a gate to which a bias voltage isapplied, and across which a bias current flows caused by the fourthcurrent source, and the output of the second amplifying circuit being ata drain of the fourteenth transistor.

This configuration can increase the gain of the first or the secondamplifying circuit by the product of the transconductance of twelfthtransistor and the output resistance thereof and by the product of thetransconductance of the fourteenth transistor and the output resistancethereof. Consequently, as the differential amplifying circuit, its gainis further increased.

As a form in another aspect, it can be configured that the firstsub-amplifying circuit of the first amplifying circuit further includes:a twenty-third transistor having a source connected to a drain of theseventeenth transistor and a gate to which a bias voltage is applied,and across which a bias current flows caused by a drain current of theseventeenth transistor; and a twenty-fourth transistor having a sourceconnected to a drain of the eighteenth transistor and a gate to which abias voltage is applied, and across which a bias current flows caused bya drain current of the eighteenth transistor, and the secondsub-amplifying circuit of the second amplifying circuit furtherincludes: a twenty-fifth transistor having a source connected to a drainof the twenty-first transistor and a gate to which a bias voltage isapplied, and across which a bias current flows caused by a drain currentof the twenty-first transistor; and a twenty-sixth transistor having asource connected to a drain of the twenty-second transistor and a gateto which a bias voltage is applied, and across which a bias currentflows caused by a drain current of the twenty-second transistor.

The gains of the first and second sub-amplifying circuits can be furtherincreased by further including the twenty-third, twenty-fourth,twenty-fifth, and twenty-sixth transistors as described above. Byincreasing the gains of the first and second sub-amplifying circuits inthis manner, the gains of the first and second amplifying circuitsincrease, and accordingly as the differential amplifying circuit, itsgain can be further increased.

Moreover, as a form, it can also be configured that the first amplifyingcircuit further includes a sixth transistor having a gate and a sourceconnected in common with those of the fourth transistor respectivelyrespectively, the second amplifying circuit further includes a tenthtransistor having a gate and a source connected in common with those ofthe eighth transistor respectively, a drain of the sixth transistor isconnected to the drain of the eighth transistor and a drain of the tenthtransistor is connected to the drain of the fourth transistor, a ratiobetween a total of gate widths converted per unit gate length of thefourth and the tenth transistor and a gate width converted per unit gatelength of the fifth transistor is nearly proportional to a current ratiobetween the first current source circuit and the second current sourcecircuit, the gate width converted per unit gate length of the fourthtransistor being equal to or more than the gate width converted per unitgate length of the tenth transistor, and a ratio between a total of gatewidths converted per unit gate length of the eighth and the sixthtransistor and a gate width converted per unit gate length of the ninthtransistor is nearly proportional to a current ratio between the thirdcurrent source circuit and the fourth current source circuit, the gatewidth converted per unit gate length of the eighth transistor beingequal to or more than the gate width converted per unit gate length ofthe sixth transistor.

This configuration can raise the parallel output resistance of thefourth and tenth transistors and the parallel output resistance of theeighth and sixth transistors, which can increase the gains of the firstand second amplifying circuits. Accordingly, as the differentialamplifying circuit, its DC gain can be increased by an increase in theDC gains of the first and second amplifying circuits.

Further, as a form, is can also be configured that the firstsub-amplifying circuit of the first amplifying circuit further includesa twenty-seventh transistor having a gate and a source connected incommon with those of the seventeenth transistor respectively, the secondsub-amplifying circuit of the second amplifying circuit further includesa twenty-eighth transistor having a gate and a source connected incommon with those of the twenty-first transistor respectively, a drainof the twenty-seventh transistor is connected to a drain of thetwenty-first transistor and a drain of the twenty-eighth transistor isconnected to a drain of the seventeenth transistor, a gate widthconverted per unit gate length of the seventeenth transistor is equal toor more than a gate width converted per unit gate length of thetwenty-eighth transistor, and a gate width converted per unit gatelength of the twenty-first transistor is equal to or more than a gatewidth converted per unit gate length of the twenty-seventh transistor.

This configuration can raise the parallel output resistance of theseventeenth and twenty-eighth transistors and the parallel outputresistance of the twenty-first and twenty-seventh transistors, which canincrease the gains of the first and second amplifying circuits. Byincreasing the gains of the first and second sub-amplifying circuits inthis manner, the gains of the first and second amplifying circuitsincreases, and accordingly as the differential amplifying circuit, itsgain can be further increased.

Based on the foregoing, the embodiments will be described below withreference to the drawings. The same numerals and symbols will be used todesignate the same or similar components in the drawings. The repeateddescription thereof will be omitted. Regarding a transistor TN (N is anumeric character or an alphabetic character), the voltage-currentconversion ratio (hereinafter expressed as transconductance) will bedescribed as gmN, and the output resistance will be described as roN.

First Embodiment

In a differential amplifying circuit 100 shown in FIG. 1, a signalvoltage IN+ inputted from a differential input terminal 1 isvoltage-to-current converted in an input stage 20 and outputted from adifferential output node 3. The outputted current is inputted to asource of an NMOS transistor T1 and outputted from a drain thereof. Thecurrent outputted from the drain of the NMOS transistor T1 is inputtedto an output stage 30 from a negative-side current output node 5 andconverted to voltage. An output signal OUT-amplified through thisprocess is outputted from a differential output terminal 7.

Similarly, a signal voltage IN-inputted from a differential inputterminal 2 is voltage-to-current converted in the input stage 20 andoutputted from a differential output node 4. The outputted current isinputted to a source of an NMOS transistor T2 and outputted from a drainthereof. The current outputted from the drain of the NMOS transistor T2is inputted to the output stage 30 from a positive-side current outputnode 6 and converted to voltage. An output signal OUT+ amplified throughthis process is outputted from a differential output terminal 8.

A gain boost amplifying circuit G1 (first amplifying circuit) isconnected to the NMOS transistor T1. An input terminal 11 of the gainboost amplifying circuit G1 is connected to the source of the NMOStransistor T1. An output terminal 13 of the gain boost amplifyingcircuit G1 is connected to a gate of the NMOS transistor T1. Similarly,an input terminal 12 of a gain boost amplifying circuit G2 (secondamplifying circuit) is connected to the source of the NMOS transistorT2. An output terminal 14 of the gain boost amplifying circuit G2 isconnected to a gate of the NMOS transistor T2. The transistor T1 and thegain boost amplifying circuit G1, and the transistor T2 and the gainboost amplifying circuit G2 constitute intermediate stages,respectively.

Referring to FIG. 2, one example of the gain boost amplifying circuitsG1 and G2 shown in FIG. 1 is described. As shown in FIG. 2, in a circuit200 as one example of the gain boost amplifying circuits G1 and G2 shownin FIG. 1, the circuit configuration is the same between the gain boostamplifying circuits G1 and G2 constituting this circuit 200. Currentsource circuits D1 and D2, an NMOS transistor T3, PMOS transistors T4 toT6, and a connection node S1 of the gain boost amplifying circuit G1correspond to current source circuits D3 and D4, an NMOS transistor T7,PMOS transistors T8 to T10, and a connection node S2 of the gain boostamplifying circuit G2, respectively.

Given a description of the gain boost amplifying circuit G1 on behalf ofthem, it includes the current source circuits D1 and D2 whose one endsare connected to a reference potential vss, the transistor T3 having asource to which the differential output node 3 is connected and a gateto which a bias voltage vb2 is applied, a current of the current sourcecircuit D1 being used as a bias current of the transistor T3, thetransistor T4 having a source connected to a reference potential vdd, adrain to which a current from the transistor T3 is inputted, and a gateconnected to a drain of the transistor T3, the transistor T5 having agate and a source connected in common with those of the transistor T4respectively and a drain to which a current from the current sourcecircuit D2 is inputted, and the transistor T6 having a gate and a sourceconnected in common with those of the transistor T4 respectively. Thesource of the transistor T3 is an input (input terminal 11) of the gainboost amplifying circuit G1, and an output (output terminal 13) of thegain boost amplifying circuit G1 is on the drain side of the transistorT5. A drain of the transistor T6 is connected to a drain of thetransistor T8, and a drain of the transistor T10 is connected to thedrain of the transistor T4.

To the gain boost amplifying circuits G1 and G2 shown in FIG. 2, PMOStransistors T11 to T14 having gates to which a predetermined biasvoltage vb1 is applied and sources to which drains of the PMOStransistor T4, T5, T8, or T9 are connected respectively are furtheradded, respectively. Given a description of the gain boost amplifyingcircuit G1 on behalf of them, it further includes the transistor T11having a source connected to the drain of the transistor T4 and a gateto which the bias voltage vb1 is applied, the current of the currentsource circuit D1 being used as a bias current of the transistor T11,and the transistor T12 having a source connected to the drain of thetransistor T5 and a gate to which the bias voltage vb1 is applied, thecurrent of the current source circuit D2 being used as the bias currentof the transistor T12. The output (output terminal 13) of the gain boostamplifying circuit G1 is at a drain of the transistor T12.

Hereinafter, the DC gain (DC voltage gain, hereinafter same as this) ofthe gain boost amplifying circuit G1 will be estimated. The DC gain ofthe gain boost amplifying circuit G1 is given by the product of a DCgain from the input terminal 11 to the connection node S1 and a DC gainfrom the connection node S1 to the output terminal 13.

First, the DC gain from the input terminal 11 to the connection node s1is estimated. The DC gain from the input terminal 11 to the connectionnode S1 is determined by the product of the transconductance of the NMOStransistor T3 and a resistance appearing at the connection node S1. Whena current ΔI is inputted from the input terminal 11, the current flowingthrough the NMOS transistor T3 also changes by ΔI. Since the biasvoltage vb2 is applied to the gate of the NMOS transistor T3, theinputted current change ΔI and a voltage change ΔVin1 of the inputterminal 11 change so as to satisfy the following equation.

ΔI=gm3*ΔVin1  (1)

The resistance appearing at the connection node S1 is estimated. If theoutput impedance of the current source circuit D1 is sufficiently large,the resistance appearing at the connection node S1 is determined by thePMOS transistors T4, T10, and T11 connected between the connection nodeS1 and the reference potential vdd. As shown in FIG. 2, the respectivedrains of the PMOS transistors T4 and T10 having sources connected tothe reference potential vdd and the source of the PMOS transistor T11having the gate to which the predetermined bias voltage vb1 is appliedare connected at a connection node S11, and the PMOS transistors T4,T10, and T11 constitute a cascode circuit.

Referring to FIG. 3, a general cascode circuit is described. In FIG. 3,a drain of an NMOS transistor T100 having a source connected to thereference potential vss and a source of an NMOS transistor T101 having agate to which a predetermined bias voltage vg is applied are connected.As is well known, a resistance Rcas of a circuit 300 appearing from adrain of the NMOS transistor T101 to the reference potential vss isroughly estimated by the product of the transconductance of the NMOStransistor T101, the output resistance thereof, and the outputresistance of the NMOS transistor T100. Namely, it is derived asfollows.

Rcas=gm101*ro101*ro100  (2)

Returning to FIG. 2, now the voltage change of the connection node S11to which the respective drains of the PMOS transistors T4 and T10 areconnected is taken as ΔVs11. Since the total of changes in currentsflowing through the PMOS transistors T4 and T10 is ΔI, the resistanceappearing between the connection node S11 and the reference potentialvdd can be written as ΔVs11/ΔI. At this time, from (2), a resistance Rs1at the connection node S1 is given as follows.

Rs1=gm11*ro11*(ΔVs11/ΔI)  (3)

Now, the resistance ΔVs11/ΔI appearing between the connection node S11and the reference potential vdd is estimated.

ΔI=ΔI1+ΔI2  (4)

where ΔI1 and ΔI2 are current changes of the PMOS transistors T4 andT10, respectively. It is assumed here that output resistances ro4 andro10 of the PMOS transistors T4 and T10 are sufficiently large. At thistime, the gate of the PMOS transistor T4 is connected to the connectionnode S1, whereby the voltage change of the transistor T4 is ΔVs1 andsatisfies the following equation.

ΔI1=gm4*ΔVs1  (5)

On the other hand, the gate of the PMOS transistor T10 is connected tothe connection node S2. Since the above operation is performed as theoperation of the differential amplifying circuit 100, when the voltageof the connection node S1 of the gain boost amplifying circuit G1changes by ΔVs1, the voltage change of the connection node S2 of thecorresponding gain boost amplifying circuit G2 becomes −ΔVs1.Consequently, the following equation is satisfied.

ΔI2=gm10*(−ΔVs1)  (6)

The following equation is obtained from (4) to (6).

ΔI=(gm4−gm10)*ΔVs1  (7)

Now, if gate widths converted per unit gate length of the PMOStransistors T4 and T10 are equal, bias voltages at the connection nodesS1 and S2 are equal, so that gm4=gm10 holds. Consequently, ΔVs11/ΔI=∞ isderived from (7), and the parallel output resistance of the PMOStransistors T4 and T10 becomes infinite. However, actually, the outputresistances of the PMOS transistors T4 and T10 are finite, which givesthe following equation:

ΔVs11/ΔI=ro4∥ro10  (8)

and the resistance ΔVs11/ΔI appearing between the connection node S11and the reference potential vdd becomes the parallel resistance of therespective output resistances of the PMOS transistors T4 and T10.

Incidentally, in order that the gain boost amplifying circuit G1functions as an inverting amplifying circuit, the sign of a resistanceΔVs1/ΔI appearing between the connection node S1 and the referencepotential vdd needs to be positive. From (7), the sign of the resistanceΔVs1/ΔI appearing between the connection node S1 and the referencepotential vdd depends on a difference gm4−gm10 between thetransconductance of the transistor T4 and the transconductance of thetransistor T10. Generally, the transconductance of a transistor isproportional to the square root of a gate width per unit gate length, sothat the gate width converted per unit gate length of the transistor T4needs to be equal to or more than the gate width converted per unit gatelength of the transistor T10. Moreover, in order to make drain voltagesof the transistors T4, T5, and T10 nearly equal, it is desirable thatthe ratio between the total of gate widths converted per unit gatelength of the transistors T4 and T10 and a gate width converted per unitlength of the transistor T5 be nearly equal to the ratio between themagnitude of the current of the current source circuit D1 and themagnitude of the current of the current source circuit D2.

From (1), (3), and (8), a voltage gain ΔVs1/ΔVin from the input terminal11 to the connection node S1 is derived as follows:

$\begin{matrix}\begin{matrix}{{\Delta \; {Vs}\; {1/\Delta}\; {Vin}} = {{gm}\; 3*{Rs}\; 1}} \\{= {{gm}\; 3*{gm}\; 11*{ro}\; 11*\left( {{ro}\; 4}||{{ro}\; 10} \right)}}\end{matrix} & (9)\end{matrix}$

and estimated by the product of the square of the transconductance andthe square of the output resistance of the transistor.

Next, the DC gain between the connection node S1 and the output terminal13 is estimated. The DC gain from the connection node S1 to the outputterminal 13 is determined by the product of the transconductance of thePMOS transistor T5 and a resistance appearing at the output terminal 13.The resistance appearing at the output terminal 13 is determined by thePMOS transistors T5 and T12 if the output resistance of the currentsource circuit D2 is sufficiently large. Since the drain of the PMOStransistor T5 having the source to which the reference potential vdd isconnected and the source of the PMOS transistor T12 having the gate towhich the predetermined bias voltage vb1 is applied are connected, thePMOS transistors T5 and T12 constitute a cascode circuit. Consequently,a resistance R13 appearing at the output terminal 13 is derived from (2)as follows.

R13=gm12*ro12*ro5  (10)

Accordingly, from (10), a DC gain ΔVout1/ΔVs1 from the connection nodeS1 to the output terminal 13 is obtained as follows:

$\begin{matrix}\begin{matrix}{{\Delta \; {Vout}\; {1/\Delta}\; {Vs}\; 1} = {{gm}\; 5*R\; 13}} \\{{= {{gm}\; 5*{gm}\; 12*r\; o\; 12*{ro}\; 5}}\;}\end{matrix} & (11)\end{matrix}$

and estimated by the product of the square of the transconductance andthe square of the output resistance of a transistor.

From the above, a DC voltage gain ΔVout1/ΔVin1 of the gain boostamplifying circuit G1 is the product of (9) and (11) and expressed asfollows:

$\begin{matrix}\begin{matrix}{{\Delta \; {Vout}\; {1/\Delta}\; {Vin}\; 1} = {\Delta \; {Vout}\; {1/\Delta}\; {Vs}\; 1*\Delta \; {Vs}\; {1/\Delta}\; {Vin}\; 1}} \\{= {{gm}\; 5*{gm}\; 12*{ro}\; 12*{ro}\; 5*{gm}\; 3*{gm}\; 11*}} \\{{{ro}\; 11*\left( {{ro}\; 4}||{{ro}\; 10} \right)}}\end{matrix} & (12)\end{matrix}$

and roughly estimated by the product of the fourth power of thetransconductance and the fourth power of the output resistance of atransistor.

Referring to FIG. 4, another example of the gain boost amplifyingcircuits G1 and G2 shown in FIG. 1 is described. In this circuit 400,the current source circuits D1 to D4 are configured as specificcircuits. In the gain boost amplifying circuit G1, NMOS transistors T31and T32 correspond to the current source circuits D1 and D2respectively, their sources being connected to the reference potentialvss, a predetermined voltage vb3 being applied to their gates, and aconstant current I being outputted from their drains. An NMOS transistorT33 has a source connected to the drain of the NMOS transistor T32, agate to which the predetermined bias voltage vb2 is applied, and a drainto which the output terminal 13 is connected. NMOS transistors T34 toT36 of the gain boost amplifying circuit G2 correspond to the NMOStransistors T31 to T33 of the gain boost amplifying circuit G1,respectively, and the NMOS transistors T34 and T35 correspond to thecurrent source circuits D3 and D4, respectively.

Resistances appearing at the connection node S1 and the output terminal13, respectively, in FIG. 4 are estimated. A resistance Rs1_4 appearingat the connection node S1 is determined by the parallel resistance of aresistance Rs1_4(vdd) appearing between the connection node S1 and thereference potential vdd and a resistance Rs1_4(vss) appearing betweenthe connection node S1 and the reference potential vss. The resistanceRs1_4(vdd) appearing between the connection node S1 and the referencepotential vdd is equal to Rs1 in (3). Since the source of the MOStransistor T3 having the gate to which the bias voltage vb2 is appliedand the drain of the NMOS transistor T31 having the source connected tothe reference potential vss are connected, the resistance Rs1_4(vss)appearing between the connection node S1 and the reference potential vssis derived from (2) as follows.

Rs1_(—)4(vss)=gm3*ro3*ro31  (13)

From (3), (8), and (13), the resistance Rs1_4 appearing at theconnection node S1 is obtained as follows.

$\begin{matrix}\begin{matrix}{{{Rs1\_}4} = \left. {{Rs1\_}4({vdd})}||{{Rs1\_}4({vss})} \right.} \\{= \left\{ {{gm}\; 11*{ro}\; 11*\left( {{ro}\; 4}||{{ro}\; 10} \right)}||{{gm}\; 3*{ro}\; 3*{ro}\; 31} \right\}}\end{matrix} & (14)\end{matrix}$

From (14), the resistance appearing at the connection node S1 is roughlyestimated by the product of the transconductance and the square of theoutput resistance of a transistor, also in the circuit in FIG. 4.

The resistance appearing at the output terminal 13 is estimated. Aresistance R13_4 appearing at the output terminal 13 is determined bythe parallel resistance of a resistance R13_4(vdd) appearing between theoutput terminal 13 and the reference potential vdd and a resistanceR13_4(vss) appearing between the output terminal 13 and the referencepotential vss. The resistance R13_4(vdd) appearing between the outputterminal 13 and the reference potential vdd is equal to R13 in equation(10). Since the source of the MOS transistor T33 having the gateconnected to the reference potential vb2 and the drain of the NMOStransistor T32 having the source connected to the reference potentialvss are connected, the resistance R13_4(vss) appearing between theoutput terminal 13 and the reference potential vss is derived from (2)as follows.

R13_(—)4(vss)=gm33*ro33*ro32  (15)

From (10) and (15), the resistance R13_4 appearing at the outputterminal 13 is obtained as follows.

$\begin{matrix}\begin{matrix}{{{R13\_}4} = \left. {{R13\_}4({vdd})}||{{R13\_}4({vss})} \right.} \\{= \left. {{gm}\; 12*{ro}\; 12*{ro}\; 5}||{{gm}\; 33*{ro}\; 33*{ro}\; 32} \right.}\end{matrix} & (16)\end{matrix}$

From (16), the resistance appearing at the output terminal 13 is roughlyestimated by the product of the transconductance and the square of theoutput resistance of a transistor, also in the circuit in FIG. 4.

From (14) and (16), the DC gain of the gain boost amplifying circuit G1shown in FIG. 4 is roughly estimated by the product of the fourth powerof the transconductance and the fourth power of the output resistance ofa transistor, similarly to the gain boost amplifying circuit G1 shown inFIG. 2.

On the other hand, the voltage gain of a differential folded-cascodegain boost amplifying circuit 1400 as a comparative reference exampleshown in FIG. 14 is estimated. The single-phase DC gain, that is, thevoltage gain from the input terminal (negative) 11 to the outputterminal (positive) 13 or the voltage gain from the input terminal(positive) 12 to the output terminal (negative) 14 corresponds to thevoltage gains of the gain boost amplifying circuits G1 and G2 in FIG. 2.

The voltage gain from the input terminal 11 to the output terminal 13 isestimated. The voltage gain from the input terminal 11 to the outputterminal 13 is the product of a transconductance gmJ7 of an NMOStransistor TJ7 of an input stage 1401 and a resistance R13_14 appearingat the output terminal 13. The resistance R13_14 appearing at the outputterminal 13 is parallel between a resistance R13_14 (vdd) appearingbetween the output terminal 13 and the reference potential vdd and aresistance R13_14(vss) appearing between the output terminal 13 and thereference potential vss. The resistance R13_14 (vdd) appearing betweenthe output terminal 13 and the reference potential vdd is determined byPMOS transistors TJ2 and TJ12 and a sub-gain boost amplifying circuitGJ1.

A source of the PMOS transistor TJ12 is connected to the referencepotential vdd. The PMOS transistor TJ2 has a source connected to a drainof the PMOS transistor T12 and a drain connected to the output terminal13. The sub-gain boost amplifying circuit GJ1 has a negative inputterminal 21 connected to the source of the PMOS transistor TJ2 and apositive output terminal 23 connected to a gate of the PMOS transistorTJ2. The transistors TJ2 and TJ12 and the sub-gain boost amplifyingcircuit GJ1 constitute an active cascode circuit.

Referring to FIG. 5, a general active cascode circuit is described. Inthis circuit 500, an NMOS transistor T200 has a source connected to thereference potential vss. An NMOS transistor T201 has a source connectedto a drain of the NMOS transistor T200. An inverting amplifying circuitG202 has a negative input terminal 203 connected to the source of theNMOS transistor T201 and a positive output terminal 204 connected to agate of the NMOS transistor T201. A resistance Rac_cas of the circuit500 appearing from a drain of the NMOS transistor T201 to the referencepotential Vss is represented by the product of the transconductance ofthe NMOS transistor T201, the output resistance thereof, the outputresistance of the NMOS transistor T200, and a DC gain Ag202 of theinverting amplifying circuit G202, and written as follows.

Rac _(—) cas=Ag202*gm201*ro210*ro200  (17)

Returning to FIG. 14, if the single-phase DC gain of the sub-gain boostamplifying circuit GJ1 is taken an Agj1, from equation (17), theresistance R13_14(vdd) appearing between the positive output terminal 13and the reference potential vdd is written as follows.

R13_(—)14(vdd)=Agj1*gmJ2*roJ12*roJ12  (18)

The resistance R13_14 (vss) appearing between the positive outputterminal 13 and the reference potential vss is determined by the NMOStransistors TJ4 and TJ10 and a sub-gain boost amplifying circuit GJ2. Asource of the NMOS transistor TJ10 is connected to the referencepotential vss. The NMOS transistor TJ4 has a source connected to a drainof the NMOS transistor TJ10. The sub-gain boost amplifying circuit GJ2has a negative input terminal 27 connected to the source of the NMOStransistor TJ4 and a positive output terminal 28 connected to a gate ofthe NMOS transistor TJ4. The transistors TJ4 and TJ10 and the sub-gainboost amplifying circuit GJ2 constitute an active cascode circuit.

Thus, if the single-phase DC gain of the sub-gain boost amplifyingcircuit GJ2 is taken an Agj2, from (17), the resistance R13_14 (vss)appearing between the positive output terminal 13 and the referencepotential vss is written as follows.

R13_(—)14(vss)=Agj2*gmJ4*roJ4*roJ10  (19)

From (18) and (19), the resistance R13_14 appearing at the positiveoutput terminal 13 is obtained as follows.

$\begin{matrix}\begin{matrix}{{{R13\_}14} = \left. {{R13\_}14({vdd})}||{{R13\_}14({vss})} \right.} \\{= \left. {{Agj}\; 1*{gmJ}\; 2*{roJ}\; 2*{roJ}\; 12}||{{Agj}\; 2*} \right.} \\{{{gmJ}\; 4*{roJ}\; 4*{ro}\; 10}}\end{matrix} & (20)\end{matrix}$

From (20), the voltage gain from the negative input terminal 11 to thepositive output terminal 13 is obtained as follows:

gmJ7(Agj1*gmJ2*roJ2*roJ12∥Ajg2*gmJ4*roJ4*roJ10)  (21)

and roughly estimated by the product of the square of thetransconductance and the square of the output resistance of a transistorand the single-phase DC gain of the sub-gain boost amplifying circuitGJ1 or GJ2.

The single-phase DC gains Agj1 and Agj2 of the sub-gain boost amplifyingcircuits GJ1 and GJ2 are estimated. The sub-gain boost amplifyingcircuits GJ1 and GJ2 can each have a fully differential folded-cascodecircuit configuration, their DC gains being equal. Hence, one example,such as shown in FIG. 15, of the gain boost amplifying circuits GJ1 andGJ2 shown in FIG. 14 is described.

The DC gain Agj1 of a fully differential folded-cascode amplifyingcircuit 1500 is a single-phase voltage gain and, for example, theproduct of the transconductance of an NMOS transistor TJS7 of an inputstage 1501 connected to the negative input terminal 21 and a resistanceappearing at an output terminal 23. A resistance R23_15 appearing at theoutput terminal 23 is parallel between a resistance R23_15 (vdd)appearing between the output terminal 23 and the reference potential vddand a resistance R23_15 (vss) appearing between the output terminal 13and the reference potential vss. Bias voltages vbj1 and vbj2 are appliedto a gate of a PMOS transistor TJS2 and a gate of an NMOS transistorTJS4, respectively, and a source of a PMOS transistor TJS12 and a sourceof an NMOS transistor TJS10 are connected to the reference potentialsvdd and vss, respectively, and therefore the following equations arederived from (10).

R23_(—)15(vdd)=gmJS2*roJS2*roJS12  (22)

R23_(—)15(vss)=gmJS4*roJS4*roJS10  (23)

From (22) and (23), the following equation is obtained.

$\begin{matrix}\begin{matrix}{{R\; 23} = \left. {{R23\_}15({vdd})}||{{R23\_}15({vss})} \right.} \\{= \left. {{gmJS}\; 2*{roJS}\; 2*{roJS}\; 12}||{{gmJS}\; 4*{roJS}\; 4*{roJS}\; 10} \right.}\end{matrix} & (24)\end{matrix}$

From (24), the single-phase DC gain Agj1 of the sub-gain boostamplifying circuit GJ1 is derived as follows:

Agj1=gmjS7*(gmJS2*roJS2*roJS12∥gmJS4*roJS4*roJS10)  (25)

and roughly estimated by the product of the square of thetransconductance and the square of the output resistance of atransistor.

From (21) and (25), the voltage gain of the gain boost amplifyingcircuit G1 shown in FIG. 14 is roughly estimated by the product of thefourth power of the transconductance and the fourth power of the outputresistance of a transistor.

From the above, it is known that the gain boost amplifying circuit G1shown in each of FIG. 2 and FIG. 4 has a DC gain equal to that of thegain boost amplifying circuit 1400 shown in FIG. 14.

Next, power consumption will be compared. The total of currents of thegain boost amplifying circuits G1 and G2 constituting the circuit 200shown in FIG. 2 is 4I since the current of each of the current sourcecircuits D1 to D4 is I. On the other hand, in the gain boost amplifyingcircuit 1400 shown in FIG. 14, if the current flowing through each ofinput transistors TJ7 and TJ8 of the input stage 1401 and transistorsTJ9 and TJ10 functioning as power sources of an output stage 1402 istaken as I, the total is 4I, which means that the same power as that ofthe circuit 200 shown in FIG. 2 is consumed by only these transistors.

Additionally, in the gain boost amplifying circuit 1400 shown in FIG.14, a transistor TJ6 is inserted between the reference potential vdd anda transistor TJ5 functioning as a current source, and thereby the amountof current which joins at the output stage 1402 via the input stage 1401from the current source transistor TJ5 is controlled by adjusting a gatevoltage vcmfb of the transistor TJ6 so that the output common modevoltage becomes the vicinity of a median value between the referencepotentials vdd and vss. The current flowing through this transistor TJ6is a current which does not directly contribute to the DC gain, andextra power is correspondingly consumed. Moreover, in the gain boostamplifying circuit 1400, the power consumption increases by an amountcorresponding to a current required for the sub-gain boost amplifyingcircuits GJ1 and GJ2.

Also in the circuit 400 shown in FIG. 4, since the current of each ofthe MOS transistors T31, T32, T34, and T35 is I, their total is 4I,which is also smaller than that in the gain boost amplifying circuit1400.

From the above, the use of the gain boost amplifying circuit 200 shownin FIG. 2 or the gain boost amplifying circuit 400 shown in FIG. 4 makesit possible to reduce power consumption as well as obtain a DC gainequal to that of the gain boost amplifying circuit 1400 as thecomparative reference example. Further, the number of required elementsreduces, so that when an integrated circuit is formed, it becomespossible to reduce the chip area and cost.

Second Embodiment

Next, another embodiment will be described with reference to FIG. 6. Asshown in FIG. 6, in a circuit 600 as still another example of the gainboost amplifying circuits G1 and G2 shown in FIG. 1, the circuitconfiguration is the same between the gain boost amplifying circuits G1(first amplifying circuit) and G2 (second amplifying circuit)constituting this circuit 600. The current source circuits D1 and D2,the NMOS transistors T3 to G6, and the connection node S1 of the gainboost amplifying circuit G1 correspond to the current source circuits D3and D4, the NMOS transistors T7 to T10, and the connection node S2,respectively.

Moreover, to the gain boost amplifying circuits G1 and G2 shown in FIG.6, the transistors T12 and T14 having gates to which the bias voltagevb2 is applied and sources respectively connected to drains of the NMOStransistors T5 and T9 are added, respectively. Given a description ofthe gain boost amplifying circuit G1 on behalf of them, it includes thetransistor T12 having the source connected to the drain of thetransistor T5 and the gate to which the bias voltage vb2 is applied, thecurrent of the current source circuit D2 being used as a bias current ofthe transistor T12. The output (output terminal 13) of the gain boostamplifying circuit G1 is at a drain of the transistor T12.

Hereinafter, the DC gain of the gain boost amplifying circuit G1 will beestimated. The DC gain from the output terminal 11 to the connectionnode S1 is the product of the transconductance of the NMOS transistor T3and a resistance appearing at the connection node S1. The resistanceappearing at the connection node S1 is determined by the NMOStransistors T3, T4, and T10. The predetermined bias voltage vb2 isapplied to a gate of the NMOS transistor T3. Sources of the NMOStransistors T4 and T10 are connected to the reference potential vss, andthe NMOS transistors T3, T4, and T10 constitute a cascode circuit.

The voltage at the connection node S1 as the gate voltage of the NMOStransistor T4 and the voltage at the connection node S2 as the gatevoltage of the NMOS transistor T10 are reverse to each other. Therefore,the DC gain from the input terminal 11 to the connection node S1 isequal to (9) and roughly estimated by the product of the square of thetransconductance and the square of the output resistance of atransistor.

The DC gain from the connection node S1 to the output terminal 13 is theproduct of the transconductance of the NMOS transistor T5 and aresistance appearing at the output terminal 13. The resistance appearingat the output terminal 13 is determined by the NMOS transistors T5 andT12. The predetermined bias voltage vb2 is applied to the gate of theNMOS transistor T12. A source of the NMOS transistor T5 is connected tothe reference potential vss, and the NMOS transistors T5 and T12constitute a cascode circuit. Therefore, the DC gain from the connectionnode S1 to the output terminal 13 is equal to (11) and roughly estimatedby the product of the square of the transconductance and the square ofthe output resistance of a transistor.

From the above, the DC gain of the gain boost amplifying circuit G1shown in FIG. 6 is roughly estimated by the product of the fourth powerof the transconductance and the fourth power of the output resistance ofa transistor, and equal to the DC gain of the gain boost amplifyingcircuit 1400 shown in FIG. 14.

Moreover, the power consumption of the gain boost amplifying circuit 600shown in FIG. 6 depends on 4I which is the total of currents flowingthrough the current source circuits D1 to D4 and smaller than that ofthe gain boost amplifying circuit 1400 shown in FIG. 14. From the above,the use of the gain boost amplifying circuit 600 of this embodimentmakes it possible to reduce power consumption as well as obtain a DCgain equal to that of the gain boost amplifying circuit 1400 shown inFIG. 14. Further, the number of required elements reduces, so that whenan integrated circuit is formed, it becomes possible to reduce the chiparea and cost.

Third Embodiment

Next, still another embodiment of the present invention will bedescribed with reference to FIG. 7. As shown in FIG. 7, in a circuit 700as yet another example of the gain boost amplifying circuits G1 and G2shown in FIG. 1, the circuit configuration is the same between the gainboost amplifying circuits G1 and G2 constituting this circuit 700.Compared with the circuit 200 shown in FIG. 2, the PMOS transistors T6and T10 are omitted, and sub-gain boost amplifying circuits GS1 and GS2are added. Now it is assumed that the sub-gain boost amplifying circuitsGS1 and GS2 have the same circuit configuration.

The DC gain of the gain boost amplifying circuit G1 shown in FIG. 7 isestimated. First, the DC gain from the input terminal 11 to theconnection node S1 is estimated. When the current ΔI is inputted to thesource of the NMOS transistor T3, the current flowing through the NMOStransistor T3 changes by ΔI. At this time, the relation between atransconductance gm3 of the NMOS transistor T3 and a voltage change ΔVinof its source is given by (1). The current change of the PMOS transistorT4 is Δ1.

The gate of the PMOS transistor T4 is connected to the connection nodeS1. Therefore, if the resistance of the current source circuit D1 issufficiently large and the transconductance of the PMOS transistor T4 istaken as gm4, a resistance ΔVs1/Δ1 appearing at the connection node S1is written as follows.

ΔVs1/ΔI=1/gm4  (26)

From (26), a voltage gain ΔVs1/ΔVin from the input terminal 11 to theconnection node S1 is obtained as follows:

ΔVs1/ΔVin=gm3/gm4  (27)

and a voltage change ΔVs1 of the connection node S1 is almost the sameas the voltage change ΔVin of the output terminal 11.

Next, the voltage gain from the connection node S1 to the outputterminal 13 will be estimated. The voltage gain from the connection nodeS1 to the output terminal 13 is the product of the transconductance ofthe PMOS transistor T5 and the resistance appearing at the outputterminal 13. If the resistance of the power source circuit D2 issufficiently large, the resistance appearing at the output terminal 13is determined by the PMOS transistors T5 and T12 and the sub-gain boostamplifying circuit GS1.

The source of the PMOS transistor T5 is connected to the referencepotential vdd. The PMOS transistor T12 has the source connected to thedrain of the PMOS transistor T5 and the drain connected to the outputterminal 13. The sub-gain boost amplifying circuit GS1 has the negativeinput terminal 21 connected to the source of the PMOS transistor T5 andthe positive output terminal 23 connected to the gate of the PMOStransistor T12. The transistors T5 and T12 and the sub-gain boostamplifying circuit GS1 constitute an active cascode circuit.Accordingly, the resistance appearing at the output terminal 13 is equalto (17) and derived as follows:

Ags1*gm12*ro12*ro5  (28)

where Ags1 is the DC gain of the sub-gain amplifying circuit GS1.

From (28), the voltage gain from the connection node S1 to the outputterminal 13 is obtained as follows:

gm5*Ags1*gm12*ro12*ro5  (29)

and roughly estimated by the product of the square of thetransconductance and the square of the output resistance of a transistorand the DC gain of the sub-gain boost amplifying circuit GS1.

The DC gains of the sub-gain boost amplifying circuits GS1 and GS2 areestimated with reference to FIG. 8. A circuit 800 shown in FIG. 8 is oneexample of the sub-gain boost amplifying circuits GS1 and GS2 shown inFIG. 7. PMOS transistors T15 and T16 in the sub-gain boost amplifyingcircuit GS1 constituting the circuit 800 have sources connected to thepositive and negative input terminals 22 and 21 respectively and gatesto which the predetermined bias voltage vb1 is applied. An NMOStransistor T17 has a source connected to the reference potential vss anda gate connected to a drain of the PMOS transistor T16. An NMOStransistor T18 has a source and a gate connected in common with those ofthe NMOS transistor T17.

Further, NMOS transistors T23 and T24 having sources connected to drainsof the NMOS transistors T17 and T18, gates to which the predeterminedbias voltage vb2 is applied, and drains connected to drains of the PMOStransistors T16 and T15, respectively.

The sub-gain boost amplifying circuits GS1 and GS2 have the same circuitconfiguration, and the PMOS transistors T15 and T16 and the NMOStransistors T17, T18, T23, and T24 of the sub-gain boost amplifyingcircuit GS1 correspond to PMOS transistors T19 and T20 and NMOStransistors T21, T22, T25, and T26 of the sub-gain boost amplifyingcircuit GS2, respectively.

The DC gain Ags1 of the sub-gain boost amplifying circuit GS1 shown inFIG. 8 is roughly estimated by the product of a DC gain from the inputterminal 21 to a connection node S21 and a DC gain from the connectionnode S21 to the output terminal 23. The DC gain from the input terminal21 to the connection node S21 is the product of a transconductance gm16of the PMOS transistor T16 and a resistance appearing at the connectionnode 21. A resistance Rs21_8 appearing at the connection node S21 isparallel between a resistance Rs21_8(vdd) appearing between theconnection node S21 and the reference potential vdd and a resistanceRs21_8(vss) appearing between the connection node S21 and the referencepotential vss. The resistance Rs21_8(vdd) appearing between theconnection node S21 and the reference potential vdd is determined by theNMOS transistor T16 and the NMOS transistor T5 connected via the inputterminal 21. Since the NMOS transistors T16 and T15 constitute a cascodecircuit, the following equation is given.

Rs21_(—)8(vdd)=gm16*ro16*ro5  (30)

Since the gate of the NMOS transistor T17 is connected to the connectionnode S21, the resistance Rs21_8 (vss) appearing between the connectionnode S21 and the reference potential vss is derived from (26) asfollows.

Rs21_(—)8(vss)=1/gm17  (31)

Thus, from (30) and (31), a rough estimate is performed as follows.

$\begin{matrix}\begin{matrix}{{{Rs21\_}8} = \left. {{Rs21\_}8({vdd})}||{{Rs21\_}8({vss})} \right.} \\{= {{1/{gm}}\; 17}}\end{matrix} & (32)\end{matrix}$

From (32), the DC gain from the input terminal 21 to the connection nodeS21 is derived as follows:

gm21/gm17  (33)

and it is small.

The DC gain from the connection node S21 to the output terminal 23 isrepresented by the product of the transconductance of the NMOStransistor T18 and a resistance appearing at the output terminal 23. Aresistance R23 appearing at the out terminal R23 is parallel between aresistance R23(vdd) appearing between the output terminal 23 and thereference potential vdd and a resistance R23(vss) appearing between theoutput terminal 23 and the reference potential vss. Since the NMOStransistors T18 and T24 clearly constitute a cascode circuit, theresistance R23 (vss) is equal to (2) and written as follows.

R23(vss)=gm24*ro24*ro18  (34)

Since the PMOS transistor T15 and the PMOS transistor T4 connectedthereto via the input terminal 22 also clearly constitute a cascodecircuit, R23(vdd) is equal to (2) and written as follows.

R23(vdd)=gm15*ro15*ro4  (35)

From (34) and (35), R23 appearing at the output terminal 23 is writtenas follows.

$\begin{matrix}\begin{matrix}{{R\; 23} = \left. {R\; 23({vdd})}||{R\; 23({vss})} \right.} \\{= \left. {{gm}\; 24*{ro}\; 24*{ro}\; 18}||{{gm}\; 15*{ro}\; 15*{ro}\; 4} \right.}\end{matrix} & (36)\end{matrix}$

From (36), the DC gain Ags1 from the connection node S21 to the outputterminal 23 is derived as follows:

Ags1=gm18*(gm24*ro24*ro18∥gm15*ro15*ro4)  (37)

and roughly estimated by the product of the square of thetransconductance and the square of the output resistance of atransistor.

FIG. 17 indicates another example of sub-gain boost amplifying circuitsGS1 and GS2 shown in FIG. 7.

As shown in FIG. 17, a circuit 850 as another example of the gain boostamplifying circuits GS1 and GS2 shown in FIG. 7 has the same circuitconfiguration as the circuit 800 shown in FIG. 8 except that a gate anda drain of PMOS transistor T15 are connected to each other, a gate and adrain of PMOS transistor T17 are connected to each other, andpredetermined bias voltage vb2 is applied to gates of NMOS transistorsT17, T18, T21, and T22.

The DC gain of the gain boost amplifying circuit GS1 shown in FIG. 17 isgiven by a product of a DC gain from the input terminal 21 to theconnection node S21 and a DC gain from the connection node S21 to theoutput terminal 23.

NMOS transistors T23 and T17 constitute current sources to provideconstant currents to the PMOS transistor T16 for stabilizing voltagebetween a gate and a source of the PMOS transistor T16. Therefore, theDC gain from the input terminal 21 to the connection node S21 is 1.

Furthermore, the DC gain from the connection node S21 to the outputterminal 23 is determined to be the product of the transconductance gm16of the PMOS transistor T15 and a resistance Rs23 appearing at theconnection node S23, as the following equation.

$\begin{matrix}\begin{matrix}{{{gm}\; 16*R\; 23} = {{gm}\; 16*\left( {R\; 23({vdd})}||{R\; 23({vss})} \right)}} \\{= {{gm}\; 16*\left( {{gm}\; 24*{ro}\; 24*{ro}\; 18}||{{gm}\; 15*{ro}\; 15*{ro}\; 4} \right)}}\end{matrix} & (38)\end{matrix}$

As the DC gain from the input terminal 21 to the connection node S21 is1, the DC gain of the gain boost amplifying circuit GS1 shown in FIG. 17is estimated to be a product of the square of the transconductance andthe square of the output resistance of the transistor. That is, the DCgains of the sub-gain boost amplifying circuits shown in FIG. 17 isequal to the DC gains of the sub-gain boost amplifying circuits shown inFIG. 8. From (37), the DC gain Ags1 of the sub-gain boost amplifyingcircuit GS1 shown in FIG. 8 is roughly estimated to be the product ofthe square of the transconductance and the square of the outputresistance of a transistor.

From (29) and (37), the gain boost amplifying circuit G1 shown in FIG. 7has a DC gain which is roughly estimated by the product of the fourthpower of the transconductance and the fourth power of the outputresistance of a transistor and equal to that of the gain boostamplifying circuit 1400 as the comparative reference example. Moreover,speaking of power consumption, it is reduced at least by an amountcorresponding to a current flowing through the transistor TJ6 of thegain boost amplifying circuit 1400. The amount of currents required inthe sub-gain boost amplifying circuits GS1 and GS2 are, for example,equal to the amount of a current required in the sub-gain boostamplifying circuit GJ1 in the gain boost amplifying circuit 1400.

Referring to FIG. 9, yet another example of the gain boost amplifyingcircuits G1 and G2 shown in FIG. 1 is described. In this circuit 900,the current source circuits D1 to D4 are configured as specificcircuits. A resistance Rs1_9 appearing at the connection node S1 isestimated. A resistance Rs1_9(vdd) appearing between the connection nodeS1 and the reference potential vdd is represented by (26). On the otherhand, since the NMOS transistors T3 and T31 constitute a cascodecircuit, a resistance Rs1_9(vss) appearing between the connection nodeS1 and the reference potential vss is equal to (2) and sufficientlylarger compared with Rs1_9(vdd). Therefore, the resistance Rs1_9appearing at the connection node S1 can approximate as follows:

Rs1_(—)9=Rs1_(—)9(vdd)∥Rs1_(—)9(vss)=Rs1_(—)9(vdd)=1/gm4  (39)

and it is equal to the resistance appearing at the connection node S1 inthe circuit shown in FIG. 7.

A resistance R13_9 appearing at the output terminal 13 is estimated.Since the PMOS resistors T5 and T12 and the sub-gain boost amplifyingcircuit GS1 constitute an active cascode circuit, a resistanceR13_9(vdd) appearing between the output terminal 13 and the referencepotential vdd is equal to (17). Since the NMOS transistors T32 and T33and the sub-gain boost amplifying circuit GS3 also constitute an activecascode circuit, a resistance R13_9(vss) appearing between the outputterminal 13 and the reference potential vdd is also equal to (17).

Referring to FIG. 10, one example of sub-gain boost amplifying circuitsGS3 and GS4 shown in FIG. 9 is described. The sub-gain boost amplifyingcircuits GS3 and GS4 constituting this circuit 1000 have the samecircuit configuration as the sub-gain boost amplifying circuits GS1 andGS2 in FIG. 8 except for the difference between PMOS and NMOS, andtransistors T15 to T26 and terminals 21 to 26 correspond to transistorsTp15 to Tp26 and terminals 21 p to 26 p, respectively. The DC gains ofthe sub-gain boost amplifying circuits GS3 and GS4 shown in FIG. 10 arethe same as the DC gain of the sub-gain boost amplifying circuit GS1since they have the same circuit configuration. If the DC gain of thesub-gain boost amplifying circuit GS3 is taken as Ags3, the resistanceR13_9 appearing at the output terminal 13 is derived as follows:

$\begin{matrix}\begin{matrix}{{{R13\_}9} = \left. {{R13\_}9({vdd})}||{{R13\_}9({vss})} \right.} \\{= \left. {{Ags}\; 1*{gum}\; 12*{ro}\; 12*{ro}\; 5}||{{Ags}\; 3*} \right.} \\{{{gm}\; 33*{ro}\; 33*{ro}\; 32}}\end{matrix} & (40)\end{matrix}$

and equal to the resistance R13 of the circuit 700 represented by (28).Hence, the DC gain of the gain boost amplifying circuit GS1 shown inFIG. 9 also becomes equal to the DC gain of the gain boost amplifyingcircuit 1400 as the comparative reference example.

FIG. 18 indicates another example of sub-gain boost amplifying circuitsGS3 and GS4 shown in FIG. 9.

As shown in FIG. 18, a circuit 1050 as another example of the gain boostamplifying circuits GS3 and GS4 shown in FIG. 9 has the same circuitconfiguration as the circuit 850 shown in FIG. 17 except that adifference between PMOS and NMOS exists. Transistors T15 to T26correspond to transistors Tp15 to Tp26, respectively. Terminals 21 to 26correspond to terminals 21 p to 26 p, respectively.

Therefore, the DC gains of the gain boost amplifying circuits GS3 andGS4 shown in FIG. 18 is equal to the DC gains of the gain boostamplifying circuits GS1 and GS2, respectively. The power consumption ofthe gain boost amplifying circuit shown in FIG. 9 is estimated. If thecurrent in each of NMOS transistors T31, T32, T34, and T35 functioningas current sources is taken as I and the current flowing through each ofthe NMOS transistors T17, T18, T21, and T22 of the sub-gain boostamplifying circuits GS1 and GS2 shown in FIG. 8 and the NMOS transistorsTp17, Tp18, Tp21, and Tp22 of the sub-gain boost amplifying circuits GS3and GS4 shown in FIG. 10 is taken as Is, the total of currents flowingthrough the gain-boost amplifying circuit 900 is derived as follows.

4I+8Is

On the other hand, the power consumption of the sub-gain boostamplifying circuits GJ1 and GJ2 used in the gain boost amplifyingcircuit 1400 is estimated. If the current flowing through each oftransistors TJS7 and TJS8 of the input stage 1501 and power sourcetransistors TJS9 and TJS10 of an output stage of the differentialamplifying circuit 1500 as a circuit diagram of the sub-gain boostamplifying circuit GJ1 is taken as Is, the current flowing through onecircuit 1500 is 4Is, and the total of currents flowing through twocircuits becomes 8Is. Accordingly, the total of currents flowing throughthe gain-boost amplifying circuit 1400 is derived as follows.

4I+8Is+(consumption current flowing through transistor TJ6)

Namely, in the gain boost amplifying circuit 1400, extra powercorresponding to the current flowing through the transistor TJ6 toadjust the output common mode voltage is consumed. Therefore, the powerconsumption of the gain boost amplifying circuits G1 and G2 used in thisembodiment is lower than that of the gain boost amplifying circuit 1400.From the above, the use of the gain boost amplifying circuits G1 and G2used in this embodiment makes it possible to reduce power consumption aswell as obtain a DC gain equal to that of the gain boost amplifyingcircuit 1400.

Fourth Embodiment

Next, yet another embodiment of the present invention will be describedwith reference to FIG. 11. As shown in FIG. 11, a circuit 1100 as yetanother example of the gain boost amplifying circuits G1 and G2 shown inFIG. 1 has the same circuit configuration as the circuit 900 shown inFIG. 9 except that the PMOS transistors T6 and T10 are added.Connections of the PMOS transistors T6 and T10 are the same as in thecase of the circuit 400 shown in FIG. 4.

The DC gain of the gain boost amplifying circuit G1 shown in FIG. 11 isestimated. A resistance appearing at the connection node S1 is given by(14) since connections of the NMOS transistors T3 and T31 and the PMOStransistors T4, T10, and T11 are equal to those shown in FIG. 4. Thus,the DC gain from the input terminal 11 to the connection node S1 becomesthe product of the square of the transconductance and the square of theoutput resistance of a transistor.

Moreover, connections between the NMOS transistors T32 and T33, the PMOStransistors T5 and T12, and the sub-gain boost amplifying circuits GS1and GS3 are equal to those shown in FIG. 9, so that the DC gain from theconnection node S1 to the output terminal 13 becomes the product of thefourth power of the transconductance and the fourth power of the outputresistance of a transistor. Consequently, the DC gain of the gain boostamplifying circuit G1 is roughly estimated by the product of the sixthpower of the transconductance and the sixth power of the outputresistance of a transistor and becomes larger than that of the gainboost amplifying circuit 1400 as the comparative reference example.

Further, since there is no current increase due to addition of the PMOStransistors T6 and T10 as estimated in the circuit 400, the totalconsumption current of the circuit 1100 is smaller by the currentflowing through the transistor TJ6 of the gain boost amplifying circuit1400, resulting in lower power consumption. From the above, the use ofthe gain boost amplifying circuits G1 and G2 used in this embodimentmakes it possible to reduce power consumption as well as obtain a DCgain larger than that of the gain boost amplifying circuit 1400.

Fifth Embodiment

Next, yet another embodiment of the present invention will be describedwith reference to FIG. 12 and FIG. 13.

In a circuit 1200 (FIG. 12) as one example of the sub-gain boostamplifying circuits GS1 and GS2 shown in FIG. 11, NMOS transistors T27and T28 are added to the sub-gain boost amplifying circuits GS1 and GS2shown in FIG. 8, respectively. The NMOS transistor T27 has a source anda gate connected in common with those of the NMOS transistor T17 and adrain connected to a drain of the NMOS transistor T21. The NMOStransistor T28 has a source and a gate connected in common with those ofthe NMOS transistor T21 and a drain connected to a drain of the NMOStransistor T17.

The DC gain of the sub-gain boost amplifying circuit GS1 shown in FIG.12 is estimated. The DC gain from the input terminal 21 to theconnection node S21 is the product of the transconductance of the PMOStransistor T16 and a resistance appearing at the connection node S21. Aresistance R21_12 (vdd) appearing between the connection node S21 andthe reference potential vdd is equal to Rs21_8(vdd) in (30). Aresistance R21_12 (vss) appearing between the connection node S21 andthe reference potential vss is determined by the NMOS transistors T17,T23, and T28.

Voltage changes of the connection node S21 of the sub-gain boostamplifying circuit GS1 and the connection node S22 of the sub-gain boostamplifying circuit GS2 are reverse because of operations as the gainboost amplifying circuits G1 and G2. Moreover, since the gate of theNMOS transistor 17 is connected to the connection node S21, and the gateof the NMOS transistor T28 is connected to the connection node S22, theconnection relation between the NMOS transistors T17 and T28 is equal tothe connection relation between the PMOS transistors T4 and T6 in thegain boost amplifying circuits GS1 and GS2 shown in FIG. 2.

Consequently, the resistance R21_12(vss) appearing between theconnection node S21 and the reference potential vss is equal to (9).From this, a resistance R21_12 appearing at the connection node S21 isequal to (14), and the DC gain from the input terminal 21 to theconnection node S21 is roughly estimated by the product of the square ofthe transconductance and the square of the output resistance of atransistor.

The DC gain from the connection node S21 to the output terminal 23 isthe product of the transconductance of the PMOS transistor T18 and aresistance R23_12 appearing at the output terminal 23 represented by(36) and roughly estimated by the product of the square of thetransconductance and the square of the output resistance of atransistor. From the above, the DC gain of the sub-gain boost amplifyingcircuit GS1 is roughly estimated by the fourth power of thetransconductance and the fourth power of the output resistance of atransistor.

A circuit 1300 shown in FIG. 13 is one example of the circuitconfiguration of the sub-gain boost amplifying circuits GS3 and GS4, andPMOS transistors T27 p and T28 p are added to the sub-gain boostamplifying circuits GS3 and GS4 shown in FIG. 8, respectively. Betweenthe circuit 1200 and the circuit 1300, their circuit configurations andconnection relations are equal except for the difference between PMOSand NMOS, and the NMOS transistors T27 and T28 correspond to the PMOStransistors T27 p and T28, respectively. Therefore, the DC gains of thesub-gain boost amplifying circuit GS3 and GS4 shown in FIG. 13 are equalto that of the sub-gain boost amplifying circuit GS1 shown in FIG. 12and roughly estimated by the product of the fourth power of thetransconductance and the fourth power of the output resistance of atransistor.

From the above, the DC gains of the sub-gain boost amplifying circuits1200 and 1300 are larger than the DC gains of the sub-gain boostamplifying circuits 600 and 1000 shown in FIG. 8 and FIG. 10.Accordingly, even where the sub-gain boost amplifying circuits 1200 and1300 are used in the gain boost amplifying circuits G1 and G2 shown inFIG. 7 and FIG. 9, a DC gain larger than that of the gain boostamplifying circuit 1400 as the comparative reference example can beobtained.

Further, since there is no current increase due to addition of the NMOStransistors T27 and T28, the total current is smaller by the currentflowing through the transistor TJ6 of the gain boost amplifying circuit1400, resulting in lower power consumption. From the above, the use ofthe sub-gain boost amplifying circuit used in this embodiment makes itpossible to obtain the gain boost amplifying circuit whose powerconsumption is reduced as well as whose DC gain is larger than that ofthe gain boost amplifying circuit 1400.

Another Embodiment

Next, one application of the differential amplifying circuit accordingto each of the embodiments described above will be described withreference to FIG. 16.

A pipelined A/D converter 1600 shown in FIG. 16 includes asample-and-hold circuit 160 which converts an inputted continuous-timeanalog signal into a discrete-time analog signal and outputs it,respective conversion stages 161, 162, . . . , 16N which each convertsthe input analog signal into a quantized signal, and a combination unit1601 which combines quantized signals outputted from the respectiveconversion stages and outputs an output digital signal.

Each of the conversion stages 161, . . . , quantizes the input analogsignal and outputs the quantized signal, and simultaneously decodes thissignal and generates a decoded analog signal. Then, it subtracts thedecoded analog signal from the input analog signal, amplifiers theresultant signal by a predetermined gain, and supplies the resultantsignal to the next conversion stage. Differential amplifying circuit A1,A2, . . . are used to subtract the decoded analog signal from the inputanalog signal and amplify the resultant signal by the predeterminedgain. Also in the sample-and-hold circuit 160, a differential amplifyingcircuit A0 is used. The differential amplifying circuit in each of theembodiments described above is suitable as these differential amplifyingcircuits A1, A2, . . . , or differential amplifying circuit A0. Namely,this can realize a low power consumption pipelined A/D convertingcircuit.

Additional advantages and modifications will readily occur to thoseskilled in the art. Therefore, the invention in its broader aspects isnot limited to the specific details and representative embodiments shownand described herein. Accordingly, various modifications may be madewithout departing from the spirit or scope of the general inventiveconcept as defined by the appended claims and their equivalents.

1. A pipeline A/D converter, comprising: a sample-and-hold unit whichconverts an inputted continuous-time analog signal into a discrete-timeanalog signal; and an A/D converter unit which converts thediscrete-time analog signal into a plurality of quantized values, theA/D converter unit having a plurality of A/D conversion stages connectedin series, the A/D conversion stage converts an inputted analog signalinto the quantized value, wherein at least one of the sample-and-holdunit and the A/D conversion stage has a differential amplifying circuitwhich includes: an input stage including a pair of differential inputterminals and a pair of differential output nodes outputtingdifferential currents according to differential voltages inputted to thepair of differential input terminals; a first intermediate stageincluding a first transistor and a first amplifying circuit, the firsttransistor having a source to which one of the pair of differentialoutput nodes and an input side of the first amplifying circuit areconnected, a gate to which an output side of the first amplifyingcircuit is connected, and a drain being a negative-side current outputnode; a second intermediate stage including a second transistor and asecond amplifying circuit, the second transistor having a source towhich another of the pair of differential output nodes and an input sideof the second amplifying circuit are connected, a gate to which anoutput side of the second amplifying circuit is connected, and a drainbeing a positive-side current output node; and an output stage using thenegative-side current output node and the positive-side current outputnode as a pair of differential input nodes and including a pair ofdifferential output terminals outputting differential voltages accordingto differential currents inputted to the pair of differential inputnodes, wherein the first amplifying circuit includes: a first and secondcurrent source circuits whose one ends are connected to a firstreference potential; a third transistor having a source to which one ofthe differential output nodes in the input stage is connected and a gateto which a bias voltage is applied, and across which a bias currentflows caused by the first current source circuit; a fourth transistorhaving a source connected to a second reference potential, a drain towhich a current is inputted from the third transistor, and a gateconnected to a drain of the third transistor; a fifth transistor havinga gate and a source connected in common with those of the fourthtransistor respectively and a drain to which a current from the secondcurrent source circuit is inputted; and a sixth transistor having a gateand a source connected in common with those of the fourth transistorrespectively, wherein the source of the third transistor is the input ofthe first amplifying circuit, and the output of the first amplifyingcircuit is on a drain side of the fifth transistor; wherein the secondamplifying circuit includes: a third and fourth current source circuitswhose one ends are connected to the first reference potential; a seventhtransistor having a source to which another of the differential outputnodes in the input stage is connected and a gate to which a bias voltageis applied, and across which a bias current flows caused by the thirdcurrent source circuit; an eighth transistor having a source connectedto the second reference potential, a drain to which a current isinputted from the seventh transistor, and a gate connected to a drain ofthe seventh transistor; a ninth transistor having a gate and a sourceconnected in common with those of the eighth transistor respectively anda drain to which a current from the fourth current source circuit isinputted; and a tenth transistor having a gate and a source connected incommon with those of the eighth transistor respectively, wherein thesource of the seventh transistor is the input of the second amplifyingcircuit, and the output of the second amplifying circuit is on a drainside of the ninth transistor; wherein a drain of the sixth transistor isconnected to the drain of the eighth transistor, and a drain of thetenth transistor is connected to the drain of the fourth transistor;wherein a ratio between a total of gate widths converted per unit gatelength of the fourth and the tenth transistor and a gate width convertedper unit gate length of the fifth transistor is nearly proportional to acurrent ratio between the first current source circuit and the secondcurrent source circuit, the gate width converted per unit gate length ofthe fourth transistor being equal to or more than the gate widthconverted per unit gate length of the tenth transistor; and wherein aratio between a total of gate widths converted per unit gate length ofthe eighth and the sixth transistor and a gate width converted per unitgate length of the ninth transistor is nearly proportional to a currentratio between the third current source circuit and the fourth currentsource circuit, the gate width converted per unit gate length of theeighth transistor being equal to or more than the gate width convertedper unit gate length of the sixth transistor.
 2. The pipeline A/Dconverter according to claim 1, wherein the first amplifying circuitfurther includes: an eleventh transistor having a source connected tothe drain of the fourth transistor and a gate to which a bias voltage isapplied, and across which a bias current flows caused by the firstcurrent source; and a twelfth transistor having a source connected tothe drain of the fifth transistor and a gate to which a bias voltage isapplied, and across which a bias current flows caused by the secondcurrent source, wherein the output of the first amplifying circuit is ata drain of the twelfth transistor, and wherein the second amplifyingcircuit further includes: a thirteenth transistor having a sourceconnected to the drain of the eighth transistor and a gate to which abias voltage is applied, and across which a bias current flows caused bythe third current source; and a fourteenth transistor having a sourceconnected to the drain of the ninth transistor and a gate to which abias voltage is applied, and across which a bias current flows caused bythe fourth current source, wherein the output of the second amplifyingcircuit is at a drain of the fourteenth transistor.
 3. The pipeline A/Dconverter according to claim 1, wherein the first amplifying circuitfurther includes a twelfth transistor having a source connected to thedrain of the fifth transistor and a gate to which a bias voltage isapplied, and across which a bias current flows caused by the secondcurrent source, and the output of the first amplifying circuit is at adrain of the twelfth transistor, and wherein the second amplifyingcircuit further includes a fourteenth transistor having a sourceconnected to the drain of the ninth transistor and a gate to which abias voltage is applied, and across which a bias current flows caused bythe fourth current source, and the output of the second amplifyingcircuit is at a drain of the fourteenth transistor.
 4. A pipeline A/Dconverter, comprising: a sample-and-hold unit which converts an inputtedcontinuous-time analog signal into a discrete-time analog signal; and anA/D converter unit which converts the discrete-time analog signal into aplurality of quantized values, the A/D converter unit having a pluralityof A/D conversion stages connected in series, the A/D conversion stageconverts an inputted analog signal into the quantized value, wherein atleast one of the sample-and-hold unit and the A/D conversion stage has adifferential amplifying circuit which includes: an input stage includinga pair of differential input terminals and a pair of differential outputnodes outputting differential currents according to differentialvoltages inputted to the pair of differential input terminals; a firstintermediate stage including a first transistor and a first amplifyingcircuit, the first transistor having a source to which one of the pairof differential output nodes and an input side of the first amplifyingcircuit are connected, a gate to which an output side of the firstamplifying circuit is connected, and a drain being a negative-sidecurrent output node; a second intermediate stage including a secondtransistor and a second amplifying circuit, the second transistor havinga source to which another of the pair of differential output nodes andan input side of the second amplifying circuit are connected, a gate towhich an output side of the second amplifying circuit is connected, anda drain being a positive-side current output node; and an output stageusing the negative-side current output node and the positive-sidecurrent output node as a pair of differential input nodes and includinga pair of differential output terminals outputting differential voltagesaccording to differential currents inputted to the pair of differentialinput nodes, wherein the first amplifying circuit includes: a first andsecond current source circuits whose one ends are connected to a firstreference potential; a third transistor having a source to which one ofthe differential output nodes in the input stage is connected and a gateto which a bias voltage is applied, and across which a bias currentflows caused by the first current source circuit; a fourth transistorhaving a source connected to a second reference potential, a drain towhich a current is inputted from the third transistor, and a gateconnected to a drain of the third transistor; a fifth transistor havinga gate and a source connected in common with those of the fourthtransistor respectively and a drain to which a current from the secondcurrent source circuit is inputted; an eleventh transistor having asource connected to the drain of the fourth transistor and a gate towhich a bias voltage is applied, and across which a bias current flowscaused by the first current source circuit; a twelfth transistor havinga source connected to the drain of the fifth transistor, and acrosswhich a bias current flows caused by the second current source circuit;and a first sub-amplifying circuit configured to perform amplificationwith the source of the eleventh transistor and the source of the twelfthtransistor as bipolar inputs and to supply an output thereof to a gateof the twelfth transistor, wherein the second amplifying circuitincludes: a third and fourth current source circuits whose one ends areconnected to the first reference potential; a seventh transistor havinga source to which another of the differential output nodes in the inputstage is connected and a gate to which a bias voltage is applied, andacross which a bias current flows caused by the third current sourcecircuit; an eighth transistor having a source connected to the secondreference potential, a drain to which a current is inputted from theseventh transistor, and a gate connected to a drain of the seventhtransistor; a ninth transistor having a gate and a source connected incommon with those of the eighth transistor respectively and a drain towhich a current from the fourth current source circuit is inputted; athirteenth transistor having a source connected to the drain of theeighth transistor and a gate to which a bias voltage is applied, andacross which a bias current flows caused by the third current source; afourteenth transistor having a source connected to the drain of theninth transistor, and across which a bias current flows caused by thethird current source; and a second sub-amplifying circuit configured toperform amplification using the source of the thirteenth transistor andthe source of the fourteenth transistor as bipolar inputs and to supplyan output thereof to a gate of the fourteenth transistor, and whereinthe first sub-amplifying circuit of the first amplifying circuitincludes: a fifteenth transistor having a source used as one of thebipolar inputs and a gate to which a bias voltage is applied; asixteenth transistor having a source used as another of the bipolarinputs and a gate to which a bias voltage is applied; a seventeenthtransistor having a source connected to the second reference potentialand a gate connected to a drain of the sixteenth transistor andoutputting a drain current to the sixteenth transistor, and aneighteenth transistor having a source and a gate connected in commonwith those of the seventeenth transistor respectively and outputting adrain current to the fifteenth transistor, and wherein the secondsub-amplifying circuit of the second amplifying circuit includes: anineteenth transistor having a source used as one of the bipolar inputsand a gate to which a bias voltage is applied; a twentieth transistorhaving a source used as another of the bipolar inputs and a gate towhich a bias voltage is applied; a twenty-first transistor having asource connected to the second reference potential and a gate connectedto a drain of the twentieth transistor and outputting a drain current tothe twentieth transistor, and a twenty-second transistor having a sourceand a gate connected in common with those of the twenty-first transistorrespectively and outputting a drain current to the nineteenthtransistor.
 5. The pipeline A/D converter according to claim 4, whereinthe first sub-amplifying circuit of the first amplifying circuit furtherincludes: a twenty-third transistor having a source connected to a drainof the seventeenth transistor and a gate to which a bias voltage isapplied, and across which a bias current flows caused by a drain currentof the seventeenth transistor; and a twenty-fourth transistor having asource connected to a drain of the eighteenth transistor and a gate towhich a bias voltage is applied, and across which a bias current flowscaused by a drain current of the eighteenth transistor, and wherein thesecond sub-amplifying circuit of the second amplifying circuit furtherincludes: a twenty-fifth transistor having a source connected to a drainof the twenty-first transistor and a gate to which a bias voltage isapplied, and across which a bias current flows caused by a drain currentof the twenty-first transistor; and a twenty-sixth transistor having asource connected to a drain of the twenty-second transistor and a gateto which a bias voltage is applied, and across which a bias currentflows caused by a drain current of the twenty-second transistor.
 6. Thepipeline A/D converter according to claim 4, wherein the firstamplifying circuit further includes a sixth transistor having a gate anda source connected in common with those of the fourth transistorrespectively; wherein the second amplifying circuit further includes atenth transistor having a gate and a source connected in common withthose of the eighth transistor respectively; wherein a drain of thesixth transistor is connected to the drain of the eighth transistor, anda drain of the tenth transistor is connected to the drain of the fourthtransistor; wherein a ratio between a total of gate widths converted perunit gate length of the fourth and the tenth transistor and a gate widthconverted per unit gate length of the fifth transistor is nearlyproportional to a current ratio between the first current source circuitand the second current source circuit, the gate width converted per unitgate length of the fourth transistor being equal to or more than thegate width converted per unit gate length of the tenth transistor; andwherein a ratio between a total of gate widths converted per unit gatelength of the eighth and the sixth transistor and a gate width convertedper unit gate length of the ninth transistor is nearly proportional to acurrent ratio between the third current source circuit and the fourthcurrent source circuit, the gate width converted per unit gate length ofthe eighth transistor being equal to or more than the gate widthconverted per unit gate length of the sixth transistor.
 7. The pipelineA/D converter according to claim 5, wherein the first amplifying circuitfurther includes a sixth transistor having a gate and a source connectedin common with those of the fourth transistor respectively; wherein thesecond amplifying circuit further includes a tenth transistor having agate and a source connected in common with those of the eighthtransistor respectively; wherein a drain of the sixth transistor isconnected to the drain of the eighth transistor, and a drain of thetenth transistor is connected to the drain of the fourth transistor;wherein a ratio between a total of gate widths converted per unit gatelength of the fourth and the tenth transistor and a gate width convertedper unit gate length of the fifth transistor is nearly proportional to acurrent ratio between the first current source circuit and the secondcurrent source circuit, the gate width converted per unit gate length ofthe fourth transistor being equal to or more than the gate widthconverted per unit gate length of the tenth transistor; and wherein aratio between a total of gate widths converted per unit gate length ofthe eighth and the sixth transistor and a gate width converted per unitgate length of the ninth transistor is nearly proportional to a currentratio between the third current source circuit and the fourth currentsource circuit, the gate width converted per unit gate length of theeighth transistor being equal to or more than the gate width convertedper unit gate length of the sixth transistor.
 8. The pipeline A/Dconverter according to claim 6, wherein the first sub-amplifying circuitof the first amplifying circuit further includes a twenty-seventhtransistor having a gate and a source connected in common with those ofthe seventeenth transistor respectively; wherein the secondsub-amplifying circuit of the second amplifying circuit further includesa twenty-eighth transistor having a gate and a source connected incommon with those of the twenty-first transistor respectively; wherein adrain of the twenty-seventh transistor is connected to a drain of thetwenty-first transistor, and a drain of the twenty-eighth transistor isconnected to a drain of the seventeenth transistor; wherein a gate widthconverted per unit gate length of the seventeenth transistor is equal toor more than a gate width converted per unit gate length of thetwenty-eighth transistor; and wherein a gate width converted per unitgate length of the twenty-first transistor is equal to or more than agate width converted per unit gate length of the twenty-seventhtransistor.
 9. The pipeline A/D converter according to claim 7, whereinthe first sub-amplifying circuit of the first amplifying circuit furtherincludes a twenty-seventh transistor having a gate and a sourceconnected in common with those of the seventeenth transistorrespectively; wherein the second sub-amplifying circuit of the secondamplifying circuit further includes a twenty-eighth transistor having agate and a source connected in common with those of the twenty-firsttransistor respectively; wherein a drain of the twenty-seventhtransistor is connected to a drain of the twenty-first transistor, and adrain of the twenty-eighth transistor is connected to a drain of theseventeenth transistor; wherein a gate width converted per unit gatelength of the seventeenth transistor is equal to or more than a gatewidth converted per unit gate length of the twenty-eighth transistor;and wherein a gate width converted per unit gate length of thetwenty-first transistor is equal to or more than a gate width convertedper unit gate length of the twenty-seventh transistor.
 10. The pipelineA/D converter according to claim 5, wherein the first sub-amplifyingcircuit of the first amplifying circuit further includes atwenty-seventh transistor having a gate and a source connected in commonwith those of the seventeenth transistor respectively; wherein thesecond sub-amplifying circuit of the second amplifying circuit furtherincludes a twenty-eighth transistor having a gate and a source connectedin common with those of the twenty-first transistor respectively;wherein a drain of the twenty-seventh transistor is connected to a drainof the twenty-first transistor, and a drain of the twenty-eighthtransistor is connected to a drain of the seventeenth transistor;wherein a gate width converted per unit gate length of the seventeenthtransistor is equal to or more than a gate width converted per unit gatelength of the twenty-eighth transistor; and wherein a gate widthconverted per unit gate length of the twenty-first transistor is equalto or more than a gate width converted per unit gate length of thetwenty-seventh transistor.
 11. The pipeline A/D converter according toclaim 4, wherein the first sub-amplifying circuit of the firstamplifying circuit further includes a twenty-seventh transistor having agate and a source connected in common with those of the seventeenthtransistor respectively; wherein the second sub-amplifying circuit ofthe second amplifying circuit further includes a twenty-eighthtransistor having a gate and a source connected in common with those ofthe twenty-first transistor respectively; wherein a drain of thetwenty-seventh transistor is connected to a drain of the twenty-firsttransistor, and a drain of the twenty-eighth transistor is connected toa drain of the seventeenth transistor; wherein a gate width convertedper unit gate length of the seventeenth transistor is equal to or morethan a gate width converted per unit gate length of the twenty-eighthtransistor; and wherein a gate width converted per unit gate length ofthe twenty-first transistor is equal to or more than a gate widthconverted per unit gate length of the twenty-seventh transistor.
 12. Thepipeline A/D converter according to claim 2, wherein the first, second,third, and fourth current source circuits each includes a transistor.13. The pipeline A/D converter according to claim 4, wherein the first,second, third, and fourth current source circuits each includes atransistor.
 14. The pipeline A/D converter according to claim 5, whereinthe first, second, third, and fourth current source circuits eachincludes a transistor.
 15. The pipeline A/D converter according to claim6, wherein the first, second, third, and fourth current source circuitseach includes a transistor.
 16. The pipeline A/D converter according toclaim 7, wherein the first, second, third, and fourth current sourcecircuits each includes a transistor.
 17. The pipeline A/D converteraccording to claim 8, wherein the first, second, third, and fourthcurrent source circuits each includes a transistor.
 18. The pipeline A/Dconverter according to claim 9, wherein the first, second, third, andfourth current source circuits each includes a transistor.
 19. Thepipeline A/D converter according to claim 10, wherein the first, second,third, and fourth current source circuits each includes a transistor.20. The pipeline A/D converter according to claim 11, wherein the first,second, third, and fourth current source circuits each includes atransistor.